1. Technical Field
The present invention relates to wireless communications and, more particularly, wideband wireless communication systems.
2. Related Art
Modern wireless radio frequency (RF) transmitters for applications, such as cellular, personal, and satellite communications, employ digital modulation schemes such as Frequency Shift Keying (FSK) and Phase Shift Keying (PSK), and variants thereof, often in combination with Code Division Multiple Access (CDMA) communication. Independent of the particular communications scheme employed, the RF transmitter output signal, sRF(t), can be represented mathematically assRF(t)=r(t)cos(2πfct+θ(t))  (1)where fc denotes the RF carrier frequency, and the signal components r(t) and θ(t) are referred to as the envelope and phase of sRF(t), respectively.
Some of the above mentioned communication schemes have constant envelope, i,e.,r(t)=R,and these are thus referred to as constant envelope communications schemes. In these communications schemes, θ(t) constitutes all of the information bearing part of the transmitted signal. Other communications schemes have envelopes that vary with time and these are thus referred to as variable envelope communications schemes. In these communications schemes, both r(t) and θ(t) constitute information bearing parts of the transmitted signal.
The most widespread standard in cellular wireless communications is currently the Global System for Mobile Communications (GSM). A second generation standard employs Gaussian Minimum Shift Keying (GMSK), which is a constant envelope binary modulation scheme allowing raw transmission at a maximum rate of 270.83 kilobits per second (Kbps). In any mobile communication system the radio spectrum is a very limited resource shared by all users. GSM employs a combination of Time Division Multiple Access (TDMA) and Frequency Division Multiple Access (FDMA) for the purpose of sharing the spectrum resource. GSM networks typically operate in the 900 MHz frequency range. The radio spectrum in the bands 890-915 MHz is for the uplink (mobile station to base station) and 935-960 MHz is for the downlink (base station to mobile station). The spectrum for both uplink and downlink is divided into 200 kHz wide carrier frequencies using FDMA, and each base station is assigned one or more carrier frequencies. Each carrier is divided into eight time slots using TDMA. Eight consecutive time slots form one TDMA frame, with a duration of 4.615 ms. A physical channel occupies one time slot within a TDMA frame. Each time slot within a frame is also referred to as a burst. TDMA frames of a particular carrier frequency are numbered, and formed in groups of 26 or 51 TDMA frames called multi-frames. While GSM is sufficient for standard voice services, future high-fidelity audio and data services demand higher data throughput rates.
General Packet Radio Service (GPRS) is a new non-voice service that allows information to be sent and received across a mobile telephone network. It supplements today's Circuit Switched Data (CSD) and Short Message Service (SMS). GPRS employs the same modulation scheme as GSM, but higher data throughput rates are achievable with GPRS since it allows for all eight time slots to be used by a mobile station at the same time.
Even higher data rates are achieved in the specification of the Enhanced Data rates for GSM Evolution (EDGE) cellular telephony standard by selectively applying a 3π/8 offset, 8-level PSK (8-PSK) modulation scheme. With this variable envelope communication scheme, the maximum bit rate is tripled compared to GSM, while the chosen pulse shaping ensures that the RF carrier bandwidth is the same as that of GSM, allowing for the reuse of the GSM frequency bands. Additionally, to further increase the flexibility of data transmission, so-called multi-slot operation has been introduced into GSM/GPRS/EDGE systems. In multi-slot operation, more than one time slot out of the eight in one GSM frame can be used for transmission with GMSK and/or 8-PSK modulation.
As mentioned above, the GMSK modulation scheme of standard GSM is an example of a constant envelope communications scheme. An example transmitter appropriate for such constant envelope modulation schemes in a mobile station unit is a translational loop transmitter. In this transmitter, the digital baseband data enters a digital processor that performs the necessary pulse shaping and modulation to some intermediate frequency (IF) carrier fIF. The resulting digital signal is converted to analog using a digital-to-analog converter (DAC) and a low pass filter (LPF) that filters out undesired digital images of the IF signal. A translational loop, essentially a phase locked loop (PLL), then translates, or up-converts, the IF signal to the desired RF frequency and a power amplifier (PA) delivers the appropriate transmit power to the antenna.
As mentioned above, the 8-PSK modulation scheme of EDGE is an example of a variable envelope communications scheme. In practice, the power spectrum emitted from an EDGE transmitter will not be ideal due to various imperfections in the RF transmitter circuitry. Thus, quality measures of the transmitter performance have been established as part of the EDGE standard and minimum requirements have been set. One quality measure that relates to the RF signal power spectrum is the so-called spectral mask. This mask represents the maximum allowable levels of the power spectrum as a function of frequency offset from the RF carrier in order for a given transmitter to qualify for EDGE certification. In other words, the spectral mask requirements limit the amount of transmitter signal leakage into other users' signal spectrum. For example, at a frequency offset of 400 kHz (0.4 MHz), the maximum allowable emission level is −54 dB relative to the carrier (dBc). Another RF transmitter quality measure of the EDGE standard is the modulation accuracy, which relates the RF transmitter modulation performance to an ideal reference signal. Modulation accuracy is related to the so-called Error Vector Magnitude (EVM), which is the magnitude of the difference between the actual transmitter output and the ideal reference signal. The error vector is, in general, a complex quantity and hence can be viewed as a vector in the complex plane. Modulation accuracy is stated in root-mean-square (RMS), 95th percentile, and peak values of the EVM and is specified as a percentage. For a given transmitter to qualify for EDGE certification, the RMS EVM must be less than 9%, the 95th percentile of EVM values must be less than 15%, and the peak EVM value must be less than 30%.
The increase in system flexibility resulting from the introduction of multi-slot operation in EDGE presents the challenge of finding an efficient implementation of a joint GMSK/8-PSK modulator which enables easy and fast switching between GMSK and 8-PSK modulation in consecutive time slots. Such modulation switching must be achieved within the so-called guard interval, merely 30 microseconds (μs) long. Further complication is encountered in the domain of the RF frequency PA. Exploiting the fact that GMSK is a constant envelope modulation scheme, the PA can typically be driven in saturation mode for higher efficiency when transmitting GSM signals. However, due to the variable envelope properties of the 8-PSK modulation option in EDGE, driving the PA in saturated mode is not possible. Rather, a certain power back-off of the PA input signal level is required to maintain adequate modulation accuracy. Typical transmitter powers may be 33 dBm in GMSK mode and 27 dBm in PSK mode. Thus, when switching modulation schemes in multi-slot operation from GMSK to 8-PSK, or vice versa, a change of PA input signal level must occur. Such change must be achieved within the guard interval and in such a fashion that switching transients do not violate the spectral mask requirements.
Another potential cause of performance degradation of RF polar transmitters is the so-called reference feed-through caused by several phase-locked loop component non-idealities, such as non-zero reset delay of the phase frequency detector (PFD) as well as mismatches between the “up” and “down” current sources of the charge pump. Reference feed-through generates tones at the RF output offset by an amount equal to the reference frequency and may lead to violation of the spurious emissions requirements of the EDGE standard. Reference feed-through may be controlled by design of the PLL signal filter. It is desirable that the PLL signal filter transfer function, H(s), equals one for all frequencies. In this case, the PLL signal filter imposes no distortion on the signal and therefore does not introduce modulation error.
In practice, designing the PLL such that H(s)=1, i.e., has infinite bandwidth, is impossible. Firstly, it can be shown that loop stability considerations dictate that the bandwidth of the PLL signal filter be less than about 1/10 of the IF frequency. Thus, for example, for a translational loop with an IF frequency of 26 MHz, H(s) must thus have bandwidth less than 2.6 MHz. Secondly, narrowing the PLL signal filter bandwidth reduces the amount of “feed-through” of the IF reference signal to the RF output signal and is thus a desirable design option. Reference feed-through is the result of several PLL component non-idealities such as non-zero reset delay of the PFD, as well as mismatches between the “up” and “down” current sources of the charge pump. These non-ideal effects create a periodic signal on the voltage controlled oscillator (VCO) control voltage corresponding to the reference frequency and are thus translated to the RF signal as spurious emission. Typically, in a high-speed digital CMOS process, the reset delay of the PFD is a few nanoseconds, and the mismatch of the charge pump current sources 5-10%.
In the prior art, the maximum narrowness of H(s) is mainly dictated by the bandwidth of the signal and the permissible modulation error. For example, in GSM, where the RMS transmitter phase modulation error performance must be better than 5° and the peak modulation error must be better than 20°, designing the PLL filter narrower than 1 MHz leads to prohibitively large modulation errors. In this case, the attenuation of reference feed-through by the PLL filter is limited and, for practical PFD reset delays and CP current source mismatches in a CMOS process, may not suffice to meet the spurious emissions requirements of the GSM standard as stated in the example.
Modulation error as a result of a narrow PLL signal filter is due to both amplitude distortion as well as group delay variation over the signal band of interest. Stated differently, group delay variation causes different frequency components of the transmitter signal to travel through the transmitter at different speeds, thereby causing inter-symbol interference. As an example, for a prior art translational loop, the modulation error resulting from the PLL signal filter is approximately 0.53° RMS. While this amount of modulation error is less than the GSM standard permits, it is typically the maximum that can be allowed in the absence of other non-ideal effects, such as analog circuit noise and non-linearities, component variations due to process variations, and component value fluctuations due to temperature variations. All of these effects add up to form the total modulation error.
Thus, in order to meet spurious emissions requirements, the translational loop RF transmitter PLL signal filter is made sufficiently narrow that worst-case reference feed-through is attenuated below the −112 dBc specification. To enable this approach, digital signal processing is employed in the baseband processor to eliminate the modulation error problems otherwise caused by a narrow PLL signal filter. Specifically, the transmit signal generated by the baseband processor is “pre-distorted” so as to counter act the distortion imposed by a narrow PLL signal filter. This “pre-distortion”, or equalization process, typically occurs in two steps: a magnitude equalizer filter pre-distorts the amplitude of the transmit signal according to the inverse of the PLL signal filter magnitude response, and a group delay equalizer filter linearizes the phase response of the entire transmitter chain, i.e., pre-distorts the transmit signal such that the combined phase response of magnitude equalizer, group delay equalizer, and PLL signal filter is linear.
However, a problem exists when the phase accumulator output increases without bound due to the quantization noise present in the equalizers. This accumulation of quantization noise gives rise to unacceptably large modulation errors. Therefore, it is clear that a need exists for a modulator that can switch between modulation modes while adhering to modulation error requirements in RF polar transmitters that are presently being designed.